Architecture for controlling a dual polarity, single inductor boost regulator used as a dual polarity supply in a hard disk drive dual stage actuator (DSA) device

ABSTRACT

A dual supply circuit uses a dual feedback control, single inductor, dual polarity boost architecture with a low side power FET for end of current recirculation sensing. A dual feedback system tracks the output voltage variations and a low side power FET end of current recirculation sensing utilizes the internal current limit sensing system. Logic defining the state of operations allows the regulator to operate in both single and dual mode to cater to wide application ranges. The positive boost regulator can be operated in a buck mode making the output voltage constant with high input supply.

BACKGROUND OF THE INVENTION

1. Field of Invention

The present invention is related to any application using a DC/DCconverter, especially when a dual polarity boost regulator is needed ina hard disk DSA. Costs related to silicon area and precision are themain concerns in integrated circuit design industry. Accordingly, thereis a continuing need to supply a high performance yet cost effectivedual polarity high voltage supplies to the DSA driver.

2. Description of Related Art

There are various ways to supply dual polarity voltage to the DSA driverfrom the low voltage supplies. A simplified schematic of a generic priorart DSA driver 10 is shown for context in FIG. 1. An input voltagesupplied from a DAC transitions from typical voltages of 0.6 volts to2.6 volts and is supplied to the positive input of amplifier 12. Aresistor feedback network is coupled between the output and the negativeinput of amplifier 12. Amplifier 12 is supplied by high voltage powersupplies. The positive power supply is +20 volts and the negative powersupply is −20 volts. The output of amplifier 12 then swings from −20volts to +20 volts, which is sufficient for powering the DSA Piezoactuator, as shown in FIG. 1.

A first prior art method and circuit 20 shown in FIG. 2 uses a negativecharge pump configuration plus a conventional boost regulator to providethe +VE and −VE power supply voltages. Control circuit 22 drivestransistor M1. Diode D1 is coupled across the current path of transistorM1. The drain of transistor M1 is coupled to node 26, which is in turncoupled to a +12 volt supply voltage through inductor L1. The positivepower supply +VE is provided by diode D4, which is coupled to node 26 atone end, and by capacitor C3, which is coupled to the other end of D4.The negative power supply −VE is provided capacitors C1 and C2, anddiodes D2 and D3. Capacitor C1 is coupled to node 26 and to the junctionof diodes D2 and D3. Capacitor C2 is coupled to diode D2 at the −VEsupply voltage node.

A second prior art method shown in FIG. 3 and FIG. 4 uses separatepositive and negative regulators 30 and 40 to provide the −VE and +VEpower supply voltages. The positive voltage is generated by an inductiveboost regulator and the negative voltage is generated by a negativecapacitor charge pump. Accordingly, FIG. 4 shows a control circuit 42for driving transistor M4 and diode D41, which are coupled to node 44.Inductor L41 is coupled between node 44 and the +12 volt supply voltage.Diode D42 is coupled between node 44 and node 46, which is the +VEsupply voltage terminal. In turn, FIG. 3 shows a control circuit 32 fordriving transistor M31 and diode D31, and transistor M32 and diode D32.The transistors are coupled between the V40 power supply and ground.Capacitor 31 is coupled between node 36, and diodes D33 and D34. DiodeD34, in turn, is coupled to capacitor C32 at node 34, which is the −VEpower supply voltage.

A third prior art method and circuit 50 shown in FIG. 5 uses a singleinductor, dual polarity architecture that is used in a number ofcommercial devices. The regulator senses the output voltages via anoperational transconductance amplifier (“OTA”) and its feedback to a MUXfor comparison with a low external low FET current limit. The high sideturns on with a fixed pulse. A control circuit 59 is used to drivetransistor M51 and diode D51, which are coupled to node 56. TransistorM52 and diode D52 are driven at node 54 and are powered by a +12 voltsupply voltage. Inductor 51 is coupled between transistor M51 andtransistor M52. Diode D53 is coupled to capacitor C51 at node 58 forproviding the +VE power supply voltage. Diode D54 is coupled tocapacitor C52 at node 52 for providing the −VE power supply voltage.

The circuit 60 shown in FIG. 6 is a more detailed version of the circuitshown in FIG. 5. Circuit 60 includes two OTAs 61 and 62 coupled tocapacitors 64 and 65 and to MUX 66. The output of MUX 66 is coupled tocomparator 68. The output of comparator 68 and a switching signal 67 areprovided to control circuit 69 for driving transistor M61 and diode D61,and transistor M62 and diode D62. Inductor L61 is coupled between nodes601 and 602. Diode D64 is coupled between node 602 and the HVP node forproviding the +VE power supply voltage. Diode D63 is coupled betweennode 601 and the HVM node for providing the −VE power supply voltage.Feedback is provided from the HVM node to amplifier 62 and resistors R61and R62. Feedback is provided from the HVP node to resistors R63 andR64. Feedback is provided from R65 to the negative input of comparator68.

In a conventional disk drive, the Voice Coil Motor (“VCM”) performs allpositioning of the head to read data located on the disk. However, withcurrent disk space demand, the track density on the disk media has growntremendously. Since this a mechanical design, the assembly of the voicecoil actuator tends to have low natural frequencies and these accumulatevibrations and cause Off-Track Errors. Therefore, one actuator is notenough to increase the data storage capacity. With the use of asecondary actuator at the tip of the main actuator, this complements thetraditional VCM actuator and forms a dual stage servo system. With manycurrent designs, this secondary actuator can be designed to have ahigher natural frequency and also less vibration. With these mechanicaldesigns, the whole system will need to be complemented with anelectrical device or drivers to drive the secondary actuator. There is,therefore, a need to generate the bias voltage for the DSA driver ofabout ±20 volts from low voltage PC supplies of +5 volts and +12 volts.

SUMMARY OF THE INVENTION

According to the present invention, a dual supply circuit uses a dualfeedback control, single inductor, dual polarity boost architecture witha low side power FET for end-of-current recirculation sensing. A dualfeedback system tracks the output voltage variations and a low sidepower FET end-of-current recirculation sensing utilizes the internalcurrent limit sensing system. Logic defining the state of operationsallows the regulator to operate in both single and dual mode to cater towide application ranges.

BRIEF DESCRIPTION OF THE DRAWINGS

The invention will be better understood by reference to the followingdescription taken in conjunction with the accompanying drawings:

FIG. 1 shows a generic prior art dual supply circuit;

FIG. 2 shows a first prior art circuit for generating dual high voltagesupplies from a low voltage power supply;

FIG. 3 and FIG. 4 together show a second prior art circuit forgenerating dual high voltage supplies from a low voltage power supply;

FIG. 5 shows a third prior art circuit for generating dual high voltagesupplies from a low voltage power supply;

FIG. 6 is a more detailed circuit schematic of the circuit shown in FIG.5;

FIG. 7 is a schematic diagram of a dual power supply circuit accordingto an embodiment of the present invention;

FIG. 8 is a simplified block diagram for a greater understanding of thedetailed circuit diagram of FIG. 7;

FIG. 9 is a partial schematic for understanding the negative boostregulator operation according to the present invention;

FIG. 10 is a partial schematic for understanding the boost regulatoronly operation according to the present invention;

FIG. 11 is a partial schematic for understanding the dual polarity boostregulator operation according to the present invention;

FIG. 12 is a set of tables for greater understanding of the logic statesdefining the operational modes of the circuit of the present invention;

FIG. 13 is a schematic diagram of a circuit used for current limitdetection and end of negative current recirculation detection accordingto the present invention;

FIG. 14 is a plot of the inductor current profile according to thepresent invention;

FIGS. 15-18 are timing diagrams showing a number of pertinent waveformsat various nodes for a greater understanding of the operational modes ofthe circuit of the present invention under high and low load conditions;and

FIG. 19 is a schematic diagram of the control logic block shown in FIG.7.

DETAILED DESCRIPTION

A circuit 70 for controlling a single inductor using a dual polarityboost regulator according to an embodiment of the present invention isshown in FIG. 7. The circuit 70 uses two voltage comparators 72 and 73to provide a dual feedback (VN and VB) to the regulator with commonreferences (VREF). This tracks the output voltage in all loadconditions. There are two end of recirculation comparators, theBST_Recir comparator 703 and the NegReg_Recir comparator 78 to senseboth inductor current decays through inductor L71 during both positiveand negative cycles of the regulator. Inductor L71 is coupled betweenthe VOUT_NEG and VOUT_BST nodes. Diode D71 is coupled to capacitor C71to provide the VNEG20V power supply voltage. Diode D72 is coupled tocapacitor C72 to provide the VBST20V positive power supply voltage.Feedback is provided from the VBST20V supply terminal through resistorsR73 and R74 to the negative input of the comparator 73. Feedback isprovided from the VBEG20V supply terminal through resistors R72 and R71to the positive input of the comparator 72.

The low side inductor current limit sensing circuit 74 is a noveldesign, integrating an internal sense FET to mirror the maximum currentof the inductor L71, and at the same time, providing an additionalinternal sense FET to mirror the current decay of the negative cycle ofthe regulator. (This is explained in further detail with respect to FIG.13, and is simplified as Ilimit sense FET 77 in FIG. 7). This mirroringof the current decay during the negative cycle of the regulator providesan end-of-recirculation signal to the main control logic, which willallow the operation states to function properly. The current limitsensing circuit also includes current sources I71 coupled to thenegative input of comparator 75 and FET 77, and I72 coupled to thepositive inputs of comparators 75 and 78.

The turning off and on of the HS FET 701 and LS FET 702 are controlledby the signals from VN, VB, Ilimit_comp, NegReg_Recir and BST_Recirprovided to control circuit 71. These signals are synchronizing eachother and turn off or turn on independently or together depending onwhat are load conditions or output voltage conditions. A detailedexplanation of the operating state is provided below. The controlcircuit drives the HS FET 702 through buffer amplifier 79, and providesthe BSTDRV signal to buffer amplifier 76, in turn driving the IlimitSense FET 77. The control block 71 referenced in FIG. 7 includesdiscrete logic that defines which mode and which states the regulatorshould operate in depending upon the input from the user and the controlsignals generated from the comparator circuits.

The BST_Recir comparator 703 is used to detect when the inductor energyis fully depleted. The BST_Recir comparator 703 detects the voltagedifference between VBST20V and VOUT_BST. When BVST20V is greater thanVOUT_BST, this means that the inductor energy is fully depleted.Comparator 78 is triggered when the inductor current flows through thelow-side FET reaches zero amps.

Turning momentarily to FIG. 19, a schematic diagram of the logic block71 is shown that provides the details of the gate level logic usedtherein. Note that the same high-side FET 701, inductor L71, low-sideFET 702, diodes D71 and D72, and capacitors C71 and C72 are used as inFIG. 7, for context. Logic block 71 includes a first logic section 71Athat includes an OR gate 760 for receiving the VB_F and VN_F signals. AnAND gate 762 is coupled to the output of OR gate 760 and receives theinverted state11 signal. AND gate 764 receives the Ilimit_comp andbst_release signals. The S input of flip-flop 766 is coupled to theoutput of gate 764 and the R input receives the Negreg_recir signal. TheS input of flip-flop 768 is coupled to the output of gate 762, the Rinput is coupled to the output gate 770. AND gate 772 receives thestate11, bst_release, and nst_release input signals. An input of NORgate 774 receives the HSFIXON signal from the Q output of flip-flop 768.The other input is coupled to the output of gate 776. The output of gate774 drives the gate of the high-side FET 701. AND gate 776 receives theinverter LSFIXON signal and the BST_DRV signal used to drive the gate oflow-side FET 702. AND gate 778 receives the Ilimit_comp and invertedLSFIXON signals. Flip-flop 780 has an S input coupled to the output ofgate 778, and an R input for receiving the BST_recir signal. Flip-flop782 has an S input for receiving the VCOMPB signal and an R inputcoupled to the Q output of flip-flop 780. NOR gate 784 has an input forreceiving the LSFIXON signal and an input coupled to the Q output offlip-flop 782. The output of gate 784 provides the BST_DRV signal fordriving low-side FET 702. AND gate 786 receives the VN_F and invertedstate11 signals. AND gate 788 receives the nst_release and state11signals. OR gate 790 has an input for receiving the VB_F signal, and aninput coupled to the output of gate 788. The S input of flip-flop 792 iscoupled to the output of gate 786, and the R input is coupled to theoutput of gate 788. The Q output of flip-flop 792 provides the LSFIXONsignal.

In FIG. 19, a second logic section 71B is also shown. In section 71B,gate 750 receives the VCOMPB and VCOMPN signals, and provides thestate11 signal. AND gate 752 receives the VB and nst_release signals,and provides the VB_F signal. The S input of flip-flop 756, and the Rinput receives the BST_recir signal. The inverted Q output provides thebst_release signal. AND gate 754 receives the VN and bst_releasesignals, and provides the VN_F signal. The S input of flip-flop 758 iscoupled to the output of gate 754, and the R input receives theNegreg_recir signal. The inverted Q output provides the nst_releasesignal.

In operation, HSFIXON path is ON for BST pumping, OFF for BST+NSTpumping, and provides PWM for NST pumping. The state11 path provides PWMfor state11 BST+NST pumping, and OFF for NST/BST pumping or for nopumping. The LSFIXON path is ON for NST pumping, and OFF for BST pumpingor for no pumping.

Turning now to FIG. 8, a block diagram 80 is provided for a greaterunderstanding of the operation of the detailed circuit 70 shown in FIG.7. Control logic 83 receives inputs 81 based upon load conditions, aswell as inputs 82 based upon user selected modes of operation. Inputs 81are the following control signals, based upon load conditions:

-   -   1. Is VOUTN>VREFN?    -   2. Is VOUTB>VREFB?    -   3. Is the ILIMIT exceeded?    -   4. Is recirculation of the negative voltage finished?    -   5. Is recirculation of the positive voltage finished?        Inputs 82 are modes selected by the user:    -   1. Boost-Buck-Boost mode.    -   2. Boost mode.    -   3. Inverting Buck-Boost mode.

Control logic 83 provides the drive signals for operating the HS switch84 and the LS switch 85, which are coupled together via an inductor L81.Diode D81 is coupled to capacitor C81 to provide the VOUT_N negativesupply voltage. Diode D82 is coupled to capacitor C82 to provide theVOUT_B positive supply voltage.

Referring back now to FIG. 7, the BST, NST, and dual polarity modes ofoperation are explained.

The Negative Boost regulator (“NST”) only operation is now explained.During a charging phase, HS FET 701 turns on and LS FET 702 turns on.Charging current (Icharging) flows through inductor L71 from the +12volt supply to ground. When maximum current has been reached, circuit 70enters into a discharging phase where HS FET 701 turns off and LS FET702 turns on. This allows recirculation of current (Irecirculation) toflow from VNEG20V to ground because the inductor current has to decay.As a result, VNEG20V goes more negative and eventually, reaching thetarget negative output voltage. In the present invention, the targetnegative output voltage is −20 volts, but of course a different targetvoltage can be chosen for a given application.

In the partial schematic 90 of FIG. 9, the Icharging and Irecirculationcurrents are shown for the NST mode of operation.

The Boost regulator (“BST”) only operation is now explained. During acharging phase, HS FET 701 turns on and LS FET 702 turns on. Chargingcurrent (Icharging) flows through inductor L71 from the +12 volt supplyto ground. During a discharging phase, when maximum current has reached,HS FET 701 turns on and LS FET 702 turns off. This allows recirculationof current (Irecirculation) to flow from +12 volts to VBST20V becausethe inductor current has to decay. As such, VBST20V goes more positiveand eventually, reaching the target positive output voltage of +20volts.

In the partial schematic 100 of FIG. 10, the Icharging andIrecirculation currents are shown for the BST mode of operation.

The dual polarity boost regulator operation is now explained. During acharging phase, HS FET 701 turns off and LS FET 702 turns on. Chargingcurrent (Icharging) flows through inductor L71 from the +12 volt supplyto ground. During a discharging phase, when maximum current has reached,HS FET 701 turn on and LS FET 702 turns off. This allows recirculationof current (Irecirculation) to flow from VNEG20V to VBST20V because theinductor current has to decay. As such, VBST20V goes more positive andeventually, reaching the target positive output voltage. Likewise,VNEG20V will go more negative and reaching target negative outputvoltage. Once recirculation of the inductor current has completed, boththe HS FET 701 and LS FET 702 will turn off.

In the partial schematic 110 of FIG. 11, the Icharging andIrecirculation currents are shown for the dual polarity boost regulatormode of operation.

Referring now to FIG. 12, the various logic states defining theoperation of circuit 70 are explained with reference to Tables A, B, andC. In Table A, two rows are shown that represent the reference voltagesfor both positive and negative boost regulator outputs at +20V and −20V,respectively. The first column is the description of the outputvoltages. The next four columns are various feedback conditions. Thesefour feedback conditions define the main four states (00, 01, 10, and11) of operation. For example in column two, if the NST does not haveenough voltage (>−20V) and BST has enough voltage (>+20V), this willdecode the “STATE 10” in Table B. For column 3, if the NST does not haveenough voltage (>−20V) and BST does not has enough voltage (<−20V), thiswill decode the “STATE 11”. In Column 4, if NST has enough voltage(<−21V) and BST has enough voltage (>+21V), this will decode the “STATE00” in. Lastly, in Column 5, if NST (<−21V) has enough voltage and BST(<20V) does not have enough voltage, this will decode the “STATE 01”.

Referring now to Table B of FIG. 12, various states are shown defined bythe output of both the voltage comparators of the tracking dualfeedbacks. Column 1 defines the state name. Column 2 defines thecomparator output name. Column 3 to column 6 defines the comparatoroutput at each operational state.

In Tables A and B of FIG. 12:

-   -   VoN=negative output voltage,    -   VoB=positive output voltage,    -   RefN=reference for negative output voltage,    -   RefB (Ref)=reference for positive output voltage,    -   column1: Both negative and positive output voltages reach output        target,    -   column2: Only negative output voltage reaches output target,    -   column3: Only positive output voltage reaches output target, and    -   column4: Both negative and positive output voltages are lower        than output.

Referring now to Table C of FIG. 12, the operation conditions of thehigh side and low side FET during each state are tabled. Column 1defines operation of the high side FET at each state, and vice versa.Column 2 defines operation of the low side FET at each state. Forexample, in “STATE 10”, (NST does not have enough voltage and BST hasenough voltage), the high side FET will be PWM and the low side FET willbe forced to turn on. Likewise for “STATE 11” (both regulators do nothave enough voltage) both high side FET and low side FET will be PWM.For “STATE 00”, both regulators have enough voltage and both high sideFET and low side FET will be off. No more pumping is required. Lastlyfor “STATE 01”—NST has enough voltage and BST does not have enoughvoltage, the high side FET will be turned on while the low side FET willbe PWM.

In Table C of FIG. 12:

-   -   HF(STATE)=operating state of high side,    -   FET, LF(STATE)=operating state of low side FET, and    -   operating state=ON/OFF/PWM (PWM=pulse width modulation).

The system is able to detect end-of-inductor current decay during “STATE10” even with low side FET fully turned on. This is achieved by a novelmaximum current detection circuit that can sense both maximum currentlimit and end of negative current decay. In addition, with respect toTables A, B, and C, VoN is the feedback voltage on the comparator forNST, and VoB is the feedback voltage on the comparator for BST. Thefeedback voltage is compared with the reference voltage using acomparator and defines its operational state.

Referring now to FIG. 13, current limit detection and end of negativecurrent recirculation detection is explained in further detail. Circuit130 is used for inductor current limit sensing. During an inductorcurrent charging phase, current flows from the supply to the LS FET andthis is mirrored with senseFET1. When the target current is developedacross the inductor, the voltage across senseFET1 rises until buffer1for senseFET1 trips and this is the maximum current limit. During theinductor current discharging phase, AND if “STATE 10”, the current willbe sensed through senseFET2. This will detect a minimum current limitwhen inductor current from NST decays to near zero. When this happens,the output voltage across senseFET2 will fall until buffer2 forsenseFET2 trips and this is the minimum current limit or negative end ofrecirculation detection. Circuit 130 includes current source I31, aswell as currents I71 and I72, referring back to FIG. 7. Transistor M131is a diode-connected transistor receiving the I131 current. TransistorM132 receives current I71, and transistor M133 receives current I72.Buffer1 corresponds to comparator 75 in FIG. 7, and buffer2 correspondsto comparator 78 in FIG. 7. The size of senseFET1, which corresponds totransistor 77 in FIG. 7 is 1/100 that of the low-side FET. The size ofsenseFET2 is one-tenth that of the low-side FET. Driver 79 from FIG. 7drives the gates of the low-side FET, as well as senseFET1, andsenseFET2. The drains of the low-side FET, senseFET1, and senseFET2, arerespectively coupled to the sources of M131, M132, and M133.

The inductor current profile is shown in the chart 140 of FIG. 14.During the charging phase 142 the inductor current reaches a maximumcurrent detect limit 144. During a discharging phase 146 the inductorcurrent reaches a minimum current detect limit 148.

Referring to the timing diagrams of FIGS. 15-18, the following signalsfrom circuit 70 in FIG. 7 are referenced:

-   -   BST_current (current associated with node VBST20V);    -   NEG_current (current associated with node VNEG20V);    -   Inductor Current (current flowing through inductor L71);    -   VN (voltage at the VN node);    -   VB (voltage at the VB node);    -   VOUT_BST (voltage at the VBST20V node); and    -   VOUT_NEG (voltage at the VNEG20V node).        Various load conditions are now presented and explained with        reference to the timing diagrams of FIGS. 15-18.

Referring now to FIG. 15, a BST High Load, NST High Load condition ispresented. When both NST and BST are below regulation voltage, theregulator enters into STATE 11. This turns on both the high side FET andthe low side FET. When both FETs are on, the inductor current starts tocharge up until the current limit has been reached. When the currentlimit has been reached, the inductor current will decay from VNEG20V toVPOS20V. As such, the VNEG20V will be more negative and VPOS20V will bemore positive. At the BST end of recirculation, the next charging cyclewill begin. Operation will continue until the next state is triggered.

Referring now to FIG. 16, a BST Low Load, NST Low Load condition ispresented. When both NST and BST have reached the regulation voltage,the regulator will be entering into “STATE00”. This will turn off bothhigh side FET and low side FET. The timing diagram of FIG. 16 shows atransition from “STATE10” to “STATE00”. The regulator will pumpoccasionally to top up the output voltage. In this case the negativeregulator output voltage is slightly not enough and the regulator willpump to top it up without charging the BST.

Referring now to FIG. 17, a BST Low Load, NST High Load condition ispresented. When NST is below regulation voltage and BST has reachedregulation voltage, the regulator will be entering into “STATE00” and“STATE10”. This will turn on low side FET and doing a PWM on high sideFET. The diagram of FIG. 17 shows a transition from “STATE10” to“STATE00”. BST has enough voltage and NST does not have enough voltage.The regulator will change the state from “STATE00” to “STATE10”. Thehigh side FET will pump and the low side FET will be turn on. Theinductor current will be charge up when high side FET until maximumcurrent limit has reached. Then high side FET will turn off untilnegative end of recirculation has detected. The state will change to“STATE00” because all voltages are good. The cycle will repeat tomaintain a constant output voltage.

Referring now to FIG. 18, a BST High Load, NST Light Load condition ispresented. When BST is below regulation voltage and NST has reached theregulation voltage, the regulator will be entering from “STATE00” and“STATE01”. This will turn on the high side FET and doing a PWM on thelow side FET. The diagram of FIG. 18 shows a transition from “STATE00”to “STATE01”. BST does not have enough voltage and NST has enoughvoltage. The low side FET will pump and the high side FET will be turnedon. The inductor current will be charged up until maximum current limithas been reached. Then the low side FET will turn off until BST end ofrecirculation has detected to start the next cycle.

The design of the present invention can operate in three modes: (1)Boost-Buck-Boost, (2) Boost, and (3) Inverting Buck-Boost. In addition,the design of the present invention has four states: (1) stopregulate—state00, (2) regulate only positive output voltage—state01, (3)regulate only negative output voltage—state10, and (4) regulate bothnegative and positive output voltage—state11. If the user chooses tooperate the regulator in a “boost-buck-boost regulator mode”, theregulator will enter and change between all of the four states aboveautomatically by load condition. If the user chooses to operate theregulator in a “boost regulator mode”, the regulator will enter andchange between states (1) and (2) automatically by load condition. Ifthe user chooses to operate the regulator in an “inverting buck-boostregulator mode”, the regulator will enter and change between states (1)and (3) automatically by load condition.

The Boost-Buck-Boost regulator mode generates higher non-inverting andinverting output voltage. The Boost regulator mode generates highernon-inverting output voltage. The Inverting Buck-Boost regulator modegenerates higher inverting output voltage. It is important to note thatthese modes are selected by the user while the four different operatingstates are defined by load conditions.

While there have been described above the principles of the presentinvention in conjunction with specific implementations of a dual powersupply circuit in accordance with the present invention, it is to beclearly understood that the foregoing description is made only by way ofexample and not as a limitation to the scope of the invention.Particularly, it is recognized that the teachings of the foregoingdisclosure will suggest other modifications to those persons skilled inthe relevant art. Such modifications may involve other features whichare already known per se and which may be used instead of or in additionto features already described herein. Although claims have beenformulated in this application to particular combinations of features,it should be understood that the scope of the disclosure herein alsoincludes any novel feature or any novel combination of featuresdisclosed either explicitly or implicitly or any generalization ormodification thereof which would be apparent to persons skilled in therelevant art, whether or not such relates to the same invention aspresently claimed in any claim and whether or not it mitigates any orall of the same technical problems as confronted by the presentinvention. The applicant hereby reserves the right to formulate newclaims to such features and/or combinations of such features during theprosecution of the present application or of any further applicationderived therefrom.

1. A single inductor dual polarity boost regulator comprising: a high side switch having an input and a first current node for receiving an input supply voltage; an inductor having a first node coupled to a second node of the high side switch; a low side switch having an input, a first current node coupled to a second node of the inductor; a first diode circuit coupled to the first node of the inductor for providing a negative output supply voltage; a second diode circuit coupled to the second node of the inductor for providing a positive output supply voltage; and a control circuit that controls the high side switch and the low side switch, the control circuit including: dual feedback controls to track the negative and positive output voltage supply variations; and a current limit detection circuit, coupled to the low side switch, for detecting a maximum current limit and an end of current decay of the inductor.
 2. The regulator of claim 1 wherein the dual feedback controls comprise two comparators sharing a single reference voltage.
 3. The regulator of claim 1 wherein the control circuit controls the high side switch and the low side switch to allow a recirculation current to flow through the inductor when the maximum current limit is detected.
 4. The regulator of claim 3 wherein the control circuit further receives a user input for configuring the regulator to be in one of a boost-buck-boost, boost, and inverting buck-boost mode.
 5. The regulator of claim 3 wherein one of the positive and negative output supply voltages can be controlled independently of the other when the regulator is in one of the boost mode and inverting buck-boost mode.
 6. The regulator of claim 1 wherein the operation state of the regulator is defined by a logic circuit coupled to a negative output comparator and a positive output comparator.
 7. The regulator of claim 1 wherein the control circuit is configured to provide a boost-buck-boost operational mode.
 8. The regulator of claim 1 wherein the control circuit is configured to provide a boost operational mode.
 9. The regulator of claim 1 wherein the control circuit is configured to provide an inverting buck-boost operational mode.
 10. The regulator of claim 1 wherein the control circuit receives a plurality of control signals based on load conditions.
 11. The regulator of claim 1 wherein the control circuit receives a plurality of control signal based on user input.
 12. The regulator of claim 1 wherein the first and second diode circuits each comprises a capacitor.
 13. A regulator circuit for driving a high side FET and a low side FET in a single inductor, dual power supply system, comprising: a control circuit for providing on, off, and PWM signals to an input of the high side FET and to an input of the low side FET; a first feedback path from a first output voltage to the control circuit; a second feedback path from a second output voltage to the control circuit; and a plurality of user control inputs to the control circuit; and a current limit detection circuit, coupled to the low side FET, for detecting a maximum current limit and an end of current decay of the inductor.
 14. The regulator circuit of claim 13 wherein the first feedback path comprises a comparator.
 15. The regulator circuit of claim 13 wherein the second feedback path comprises a comparator.
 16. The regulator circuit of claim 13 further comprising first and circuit recirculation current comparators.
 17. A regulator circuit for driving a high side FET and a low side FET in a single inductor, dual power supply system, comprising: a control circuit for providing on, off, and PWM signals to an input of the high side FET and to an input of the low side FET; a first feedback path from a first output voltage to the control circuit; a second feedback path from a second output voltage to the control circuit; and a current limit circuit coupled to the low side FET for detecting a maximum current limit and an end of current decay of the inductor.
 18. The regulator circuit of claim 17 wherein the current limit circuit comprises: a first sense FET having a gate coupled to a gate of the low side FET; and a second sense FET having a gate coupled to the gate of the low side FET.
 19. The regulator circuit of claim 17 further comprising a first FET coupled to a drain of the low side FET, a second FET coupled to a drain of the first sense FET, and a third FET coupled to a drain of the second sense FET.
 20. The regulator circuit of claim 19 further comprising first, second, and third current sources coupled to the first, second, and third FETs. 